Photodiode bootstrap phase rolloff

Hi all,
I am currently working on a range-switched photodiode design for use in
our laboratory [1]. It took me a while, but I've arrived at a design I'm
fairly happy with (in no small part thanks to Phil Hobbs' helpful
writings on the topic):
My current draft uses a simple BF862 source follower at the summing node
of a ADA4817-based 2 M?/2 k? transimpedance amplifier to bootstrap the
40 pF of photodiode capacitance (its output being AC-coupled into the PD
bias node). On paper and in SPICE, I am getting just over 2 MHz
bandwidth on the 2 M? range at reasonable noise levels (limited by the
single BF862's e_n), assuming fairly realistic parasitics. On the lower
gain range, however, I'm only predicting ~75 MHz of bandwidth, as the
BF862 stage quickly loses steam driving the 40 pF load due to its output
impedance.
This gets better when putting an active sink on the JFET source, but I
thought I would spend some time optimizing the bootstrap design while
I'm waiting for parts to arrive (~100 MHz bandwidth would be nice to
have). It's not hard to come up with a circuit that provides >0.99 gain
into the given load out to a few hundred MHz in simulation, for example
by adding an RF(-ish) PNP transistor to make a complementary feedback
pair, or simply tacking on an NPN emitter follower (bootstrapping the
JFET drain for better effective gain as necessary).
However, while those bootstrap designs look good in isolation, it seems
like they invariably render the complete circuit unstable on the 2 k?
range in simulation (although they work brilliantly on the 2 M? range).
I'll have to go through the maths properly, but this appears to be due
to the two-pole rolloff caused by having two transistors in the path. On
the 2 M? range this happens far above the op-amp loop bandwidth anyway,
but this is no longer true on the 2 k? range. The bare BF862, on the
other hand, has a lower gain to start with and rolls off earlier, but
the single pole doesn't seem to degrade the phase margin enough to cause
issues.
Building a sub-nV emitter follower that is good out to multiple GHz
obviously can't be the right solution. I feel like I'm just missing an
easy fix in the form of a lead-lag-type tweak in the right place, but I
can't quite figure out where that would be, and much less how to make
that work without degrading the noise performance too much.
Since my intuition for analog design is not great, I thought I'd ask
here whether I missed something obvious. I can also post a sketch or
LTspice file of the relevant part of the circuit if that helps, but I
figured the question would be trivial to answer for the photodetector
veterans among you, as presumably you would have run into this before.
Thanks!
? David
[1] Quantum optics. The idea is to build small (well, small as far as
lab gear goes ? ~ 35mm x 30mm x 25mm) modules with built-in beam
sampling optics for laser intensity monitoring and stabilisation all
across the visible and near-IR. And yes, the range switching led to some
rather expendable gymnastics, but the alternatives would have been even
more painful.
Reply to
David Nadlinger
Loading thread data ...
rap the
he
Sounds nice. (2MHz at 2 Meg ohm.) I'm only a tadpole when it comes to PD circuits, so I can't help much. I'm guessing a schematic would help those with more experience. How are you switching the gain? Maybe three orders of magnitude is too much. Could you use a second ADA4817 to bootstrap the PD? (I'm mostly a brute force type.)
George H.
?
e).
ay,
Reply to
George Herold
m
e
strap the
D
the
t
Oh and what photodiode? How much is the reverse bias? Though I not looked at how much the capacitance changes, you can reverse bias PD's much more than the the typical spec sheet claims.... that might gain you a little speed. GH
?
nge).
n
yway,
he
e
m
e
Reply to
George Herold
One approach is to switch the whole front end and not just the feedback res istor.
The BF862 is slower than the op amp. I've had good luck with a BFT92 PNP wr aparound (shunt feedback) on the FET up to about 100 MHz.
For faster stuff I like cascoding an ATF38143 pHEMT with a BFP640 SiGe:C bi polar. As a follower you'd need to bootstrap the drain, i.e. AC couple the BJT's base to the pHEMT's source. Use a BLM18BB-series 5- or 10-ohm bead i n series with the base to keep the BJT from oscillating.
The result is a pretty good follower out to a gigahertz or so.
Cheers
Phil Hobbs
Reply to
pcdhobbs
If reality doesn't differ from the data sheets too much, it is going to be a Hamamatsu S6775; ~5 mm x 5 mm active area with 40 pF capacitance at -10 V bias.
The PD package itself seems to be rather low capacitance, so I could indeed go to voltages higher than the 12-15 V I was going to use, and hence increase the "range" of the simple BF862 bootstrap. There are two drawbacks to this, though:
First, the dark current more than doubles going from 10 V to 30 V according to the datasheet (still waiting for the diodes to arrive to verify this). The shot noise from that still wouldn't be an issue, but it looks like the dark current will be the largest source of temperature drifts in the system.
(One important use of the photodiodes will be for intensity stabilisation, although the fact that there is a non-negligible amount of free space optics afterwards makes it hard to get better than 0.1% long-term stability anyway.)
Secondly, I would need an appropriate bias voltage source that can supply ~1 mA of current (full range at 2 k? would be ~2 mA, but hopefully no more than 0.5 mA of photocurrent will ever be used in practice). Since the PCB is going to be rather small (~ 30mm x 20mm), appropriately shielding a boost converter seems a bit nontrivial, so I would probably have to supply this externally.
Also, since there are going to be quite a few of the photodiodes in the experiment, I'd rather not push any parts past their documented limits without good reason, as I don't want to end up having to constantly fix the hardware.
Thanks for the suggestion, though!
? David
Reply to
David Nadlinger
It certainly looks like it will turn out nicer than what you can buy off the people who bribe physicists with red snack boxes, yes. 1 MHz would also be acceptable for the application, so I'm quite optimistic that I'll be able to make it work.
I'm using three semiconductor switches in a T configuration to switch the low-impedance feedback path in and out (the middle leg dragging the trace to ground when not used to reduce feedthrough). Getting the parasitics low enough to not degrade the 2 M? range was a bit of a challenge, but it turns out to be just about achievable without using (comparatively) huge relays.
Perhaps I should have made this clearer in the original post, but it's not actually the range switching that is the problem. I merely mentioned it to provide some background for the design choices I made, as some of the tradeoffs might be different if I only had a single 2 k? range to worry about.
I could, but this would increase the system noise by a factor of about five ? the ~0.8 nV/rtHz voltage noise of the BF862 across the 40 pF photodiode capacitance already dominates the design, so the 4 nV/rtHz of the ADA4817 look quite bad. The input capacitance of the op-amp is also higher than that of a BF862 source follower, let alone a bootstrapped one.
? David
Reply to
David Nadlinger
OK, I've never looked at the dark current closely. I was thinking 40-50 volts, making it quite is a bit of work. (maybe some batteries? I know, I don't really like batteries.)
0.1% ! I often find some fingerprint/ dust spec, is right in the beam making all sorts of bouncy noise. But don't you want to mostly take out the high frequency noise? (Well that's why I need a faster PD anyway.)
George H.
Reply to
George Herold
o
Oh I was asking for my own edification. I've been recently trying to speed up my PD and the switch/stray/ capacitance (grayhill rotary switch) is enough such that I need no compensating cap. at 100k ohm with a 3dB point of ~800kHz (? data book is at work).. big PD ~130 pF @10V. ..
pF
.
Huh, (Thanks) I was thinking it'd only double the noise of the opamp. I hadn't thought about the bootstrap cutting down the v_noise*C_in term, but only about more speed.
That does mean (I think) that if I bootstrap my circuit with an opamp it'll only increase the noise a bit.
4nV/rtHz is not too bad... (my opamp is much worse, 8nV.. four times as bad in my thinking... noise should be expressed as a density. V^2/Hz and not these weird EE units. :^)
George H.
George h.
Reply to
George Herold
You want to add an emitter-follower to the BF862, to lower Zout and isolate it from the PD capacitance its driving. You'll want a low e_n for that transistor. For example, I use an FMMT718 running at 2 mA. I measure a 45MHz bandwidth for the bootstrap.
Please check in AoE III, chapter 8, for details on all of the above, valuable e_n measurements, and other TIA amp tricks.
BTW, I don't agree with Phil concerning these TIA bootstraps; one does fine for any gain over 0.95 or so, because you're reducing the PD's effective capacitance by 10 to 20x, and thus its en-C effects against the more noisy op-amp, which is already probably more than you need.
--
 Thanks, 
    - Win
Reply to
Winfield Hill
It's a lot easier to get a smooth, predictable transfer function and low phase whoopdedoos with a super good bootstrap, though, particularly with high capacitance. Otherwise you have to try cancelling the residual RC pole in the second stage, which generally works OK except that the cancellation is never perfect and hence you get late-time settling artifacts.
Plus I often like to use high slew (and thus noisy) amps such as the LM6171 as the TIA because it greatly reduces artifacts due to slew limiting. In my designs the first couple of stages are much faster than the overall TIA, because you never know what sort of nasty sharp pulses the customer is going to send into the photodiode.
Cheers
Phil Hobbs
--
Dr Philip C D Hobbs 
Principal Consultant 
 Click to see the full signature
Reply to
Phil Hobbs
One other thing: I much prefer the PNP wraparound trick to adding an emitter follower, because it doesn't degrade the noise the same way.
Cheers
Phil Hobbs
--
Dr Philip C D Hobbs 
Principal Consultant 
 Click to see the full signature
Reply to
Phil Hobbs
Hi Phil (and others),
First of all, thank you very much for your message, and sorry for this late reply ? I was traveling abroad and had trouble accessing my news server.
I initially considered switching the entire frontend as a way of tolerating more parasitic capacitance (vs. switching the TIA feedback impedance). But after figuring out a workable configuration for that, it seemed like sharing the frontend would be the less complex option (which is handy given the small target PCB size).
I suppose splitting up the frontends would allow me to trade off voltage noise vs. current noise on the lower gain range, though, or go for a higher 1/f knee but lower wideband noise, and would also make it easier to add in an "ideal" output lowpass filter.
FET up to about 100 MHz.
Yep, this is what I was (probably in error) referring to as a complementary feedback pair. In fact I was even going to try the BFT92, as I have some BF{R,T}92s lying around. (They seem to be a good starting point for low-noise, medium-speed things like this ? any other favourites I should be aware of?)
[?]
Okay, this part was supposed to be a further explanation of how I couldn't get the loop stable in SPICE once I put the follower or wraparound in. However, I had only tried follower designs that were stable (and didn't peak beyond unity gain) in isolation before. Just going with a "bare" PNP wraparound without any capacitors to roll off the loop gain seems to work nicely in the full circuit in SPICE. Even with a simple resistor load in the source, the performance looks to be adequate. (For posterity, the circuit in SPICE:
formatting link

Bootstrapping the drain with a BFR92A from a BFR92A emitter follower and using active current sinks it looks like I could push the design past 200 MHz with 40 pF of input capacitance, but that's entirely unnecessary for the application.
I'll have to have a closer look at the biasing given device variations and the drift performance once I've built the thing, but due to a mess-up in our department's finance office I've been waiting for a Digi-Key order to arrive for more than a month now?
Although overkill for this application, I'll definitely keep that one in my bag of tricks. I presume you prefer the BLM18BB beads because of their wide resistive region to higher frequencies? I'm also not sure I understand how to build a unity gain follower from it, but I'll look into it more closely some other day.
Cheers!
? David
Reply to
David Nadlinger
Here's a circuit that switches a photodiode between two amps. It uses's Phil's bootstrap cascode arrangement, and switches two cascode transistors to steer the pd current.
formatting link

This worked fine; the real problem was the customer.
formatting link

--
John Larkin         Highland Technology, Inc 

lunatic fringe electronics
 Click to see the full signature
Reply to
John Larkin
news
k resistor.
P wraparound (shunt feedback) on the
:C bipolar. As a follower you'd need to bootstrap the drain, i.e. AC coupl e the BJT's base to the pHEMT's source. Use a BLM18BB-series 5- or 10-ohm b ead in series with the base to keep the BJT from oscillating.
That's nice, Thanks. The stuff down the bottom (labeled comp.) is to take care of the bias current of the cascode?
George H.
Reply to
George Herold
Nifty. Are AD8034s OK driving 10 uf like that, though?
I had one customer who used a setup like U3A/B to share a low- noise reference with several ADCs. It oscillated big-time, but they actually got away with it because the effect was common to all of the channels.
-- john, KE5FX
Reply to
John Miles, KE5FX
R12 bleeds a little current in to the cascode transistor to keep it on, low emitter impedance, at low photodiode currents. That creates a DC offset error. R6 and Q1 create a simulated equivalent current that is used to cancel the offset. Cute but overkill maybe.
--
John Larkin         Highland Technology, Inc 
picosecond timing   precision measurement  
 Click to see the full signature
Reply to
John Larkin
One of my hobbies is characterizing opamps for c-load behavior. This one is OK.
Many RRO opamps are c-load stable with a big C.
--
John Larkin         Highland Technology, Inc 
picosecond timing   precision measurement  
 Click to see the full signature
Reply to
John Larkin

ElectronDepot website is not affiliated with any of the manufacturers or service providers discussed here. All logos and trade names are the property of their respective owners.