It's always fun to compare microstrip calculators, especially at high w/h ratios.
Saturn only allows w/h up to 3, so I tried that. W=30 mils, T=10, Er=4.6.
Saturn 29.42 ohms
Appcad 35.70 ohms
Txline 35.71 ohms
Saturn may be using one of the old classical microstrip formulas. They get bad at high w/h, and many go negative. There are some really bad javascript ones online.
-- John Larkin Highland Technology Inc www.highlandtechnology.com jlarkin at highlandtechnology dot com Precision electronic instrumentation
Methinks you might be referring to mcalc, which the author admits has problems and was replaced by wcalc: Windoze installer at: Note the front ends for Octave, Matlab, Scilab, etc.
Using your example, I get 35.03 ohms Looks like Saturn may have a problem. Thanks.
-- Jeff Liebermann jeffl@cruzio.com 150 Felker St #D http://www.LearnByDestroying.com Santa Cruz CA 95060 http://802.11junk.com Skype: JeffLiebermann AE6KS 831-336-2558
Actually, I was referring to the type of formula that was in the Motorola ECL handbook, and in similar semi appnotes.
Any single formula for microstrip impedance will be valid over a limited range of geometries. Wide traces made negative impedances in the Moto formula.
-- John Larkin Highland Technology, Inc jlarkin att highlandtechnology dott com http://www.highlandtechnology.com
interesting
I tried old Dimstrip from the DOS days and got 36.34 ohms, remember the RF/microwave engineers used to claim 1% accurate?
then went to a crude femm analysis at 1MHz and got 37.94 we're talking a 6% error, but what's correct? I'd go to the PCB ouses
With ordinary FR4, epsilon can range from about 3.8 to 4.5, so your trace impedances are going to be uncertain by about
deltaZ/Z ~ 1-sqrt((1+3.8)/(1+4.5)) ~ 7%
anyway. That's another reason to ask the board house.
Cheers
Phil Hobbs
-- Dr Philip C D Hobbs Principal Consultant ElectroOptical Innovations LLC Optics, Electro-optics, Photonics, Analog Electronics 160 North State Road #203 Briarcliff Manor NY 10510 hobbs at electrooptical dot net http://electrooptical.net
On Wednesday, April 16, 2014 9:43:34 AM UTC-7, Phil Hobbs wrote: [on modelling transmission lines in printed wiring]
And, it's another reason to include a test coupon! The board house SHOULD know, but they're certainly as reliable as the watchful eye of their customers demands...
Board houses also want some tolerance on the dielectric thickness.
You can pay them to target the actual trace impedances, but that can get complex and expensive.
I usually hit about +-10% with reasonable tolerances and just asking for FR4. That's good enough for most applications. It's best to try to design so that it doesn't matter much. That's easy for digital boards.
-- John Larkin Highland Technology, Inc jlarkin att highlandtechnology dott com http://www.highlandtechnology.com
I often include a couple of SMA footprints on a board, and connect them with a 50 ohm trace that hops through all the layers. I can install the connectors and TDR that and see how well I did on trace impedances.
You can see the connector transition and the vias, too. The 30 ps scope rise time gets trashed by the trace losses.
-- John Larkin Highland Technology, Inc jlarkin att highlandtechnology dott com http://www.highlandtechnology.com
The Hayward book is out of print, btw. Good book.
I still wonder why you can't filter them out. This is usually "easy," so this naturally leads to "why not?"
Has some treatment of noise: Fundamentals of Mixer Design, Agilent, Steve Long
other mixer stuff
----------------- Well known papers, but didn't find online: Sources of intermodulation in diode-ring mixers, H. P. WALKER Predicting Intermodulation Suppression in Double-Balanced Mixers, Bert Henderson, WJ Tech Note The Relationship Between Cross-Modulation and Intermodulation Distortions in the Double-balanced Modulator, JOHN G. GARDINER
(The Walker paper is cited a great deal.)
Easy hits:
Nice. Thanks.
The carrier frequency is NOT fixed, and could go low enough that over 50 harmonics are lower than that first fundamental. Not easy to filter that out. Tracking filters? can I get more thsn 60dB rejection to the third? I'd like 80 dB. Any upper harmonic slipping in causes problems. Perhaps, tracking filter will be equivalent. I like the idea of using hard driven diode ring mixers for their simplicity. just have to add that elusive tracking filter to rid of anything much above the fundamental. If this were audio, I see a way to do it, but then again if audio, wouldn't have to, multipliers exist.
Thank you! Now I know what it feels like to take a drink from a firehose!
What do you work in? TI email address? does TI make a high speed analog multiplier? Multiplier would help me. Something with carrier to 100MHz, or
200MHz, or 500MHz Harris (Intersil) seems to make a ADL???? 1.2GHz multiplier, that looks like it has 60dB rejection up to 100MHz but that's the dreaded typical spec, which means: who knows what?
Why do you insist of generate the modulated carrier at the final frequency ?
Generate a fixed, say 300 MHz carrier, modulate it, run resolution through a fixed 300 MHz filter with required bandwidth (typically twice the modulating frequency). Mix this with a LO (VFO, PLL, DSS) variable between 300.01 to 400 MHz, remove the image frequency (600-700 MHz) with a simple LPF at 120 MHz.
Analog audio sweep generators for the 20 Hz to 20 kHz range in a single sweep with linear sweep (xx Hz/s) has been used for at least 50 years. These generated a narrow band sweep at say 100-120 kHz and was then mixed down to 0..20 kHz,followed by an LPF at say 30 kHz. This is the opposite of how general coverage HF receivers (0..30 MHz) have been operated for a few decades. The RF signal is mixed up to 45 MHz with the LO tuning between 45..75 MHz. At the 45 MHz IF the first filter has a bandwidth of about 25 kHz, which is then mixed down to
10.7 MHz or 455 kHz for NBFM.Spectrum analyzers usually work also in this upconverter mode, with the first IF somewhere in 1..3 GHz.
Thanks!!! Your description of the architecture jogged my mind enough to envision a way to do this!
Now it's off to Minicircuits for parts. ;)
When using LO high side mixing, please remember to switch side band in any SSB type transmission.
With LO high side mixing topologies, you also need to consider the frequency accuracy and phase noise issues of the LO.
A 1 ppm frequency error at final carrier of 1 MHz is 1 Hz away. However, in an upconverting system, in which the LO runs in 300-400 MHz range and in this case at 301 MHz, 1 ppm is 300 Hz frequency error.
I have seen uncalibrated audio signal generators set to -100 Hz in order to get a decent +20 Hz signal :-).
These days with NCO/DDS systemic running at several hundred MHz, this should no longer be an issue.
Well, I think Mr Upside Down Under has given you the jist of the idea that I was more or less assuming (and that is why I didn't get it). Maybe that will work for you. Enjoy the spur charts!
I am in T&M these days, and on the RX'er side. (Spec An, Signal monitoring)
The TI email address is long gone. I was there 2001 to late 2004, after Metricom's crash. I keep the google address because it is a dead end for spammers.
Don't know. Don't think so. They spun off the WLAN stuff, which is where I was located.
After your excellent suggestion to become aggressive on the frequency shifting, I blocked out the system into 'physically realizable' RF components [I think} Once into that form, it was easy to see 'weaknesses', the sensitivity to phase noise, etc. So far, everything is under 500MHz, which should make for easily obtainable components. One thing that came out, was that the MOST sensitive transition can be made to be fixed, which means it's possible to do a bit of tuning to enhance performance.
Thanks again. My only defense for NOT pursuing this architecture earlier is that I greatly feared getting into those noise sources - that's my story, and I'm sticking to it.
If you derive all the LOs from the same good quality crystal oscillator, _and_keep_the_path_delays_the_same_, most of that added phase junk will get subtracted again on the way down.
Cheers
Phil Hobbs
-- Dr Philip C D Hobbs Principal Consultant ElectroOptical Innovations LLC Optics, Electro-optics, Photonics, Analog Electronics 160 North State Road #203 Briarcliff Manor NY 10510 hobbs at electrooptical dot net http://electrooptical.net
If you are using a DDS for frequency generation, some chips have a DDS clock frequency premultiplier to allow use of fundamental mode crystals (below 20 MHz) directly. These premultipliers are PLLs, which might have an effect on phase noise, so check the specs with or without premultiplier.
Since you apparently have only two frequencies f1 and f2 of interest, you could even use a spreadsheet to validate your frequency allocations and the locations of any spurs.
Put integer multiples (both positive and negative) horizontally and multiples of f2 vertically on the spread sheet. In each cell, calculate the sum of the column header and row header. In one cell, there will be the wanted signal f1-f2 and in other cells various unwanted frequencies.
To get a better visual overview, take the absolute value of the sum (to reflect the negative sums) and compare the result the LPF bandwidth, in your case somewhere between 120 to 150 MHz) and anything above this can be filtered out and the cell should also be set to spaces, instead of any numeric values, thus, any offending frequencies will shown clearly.
Since the amplitude of the spurs seem to drop rapidly with higher intermodulation products, you should not have to care e.g. for -11 x f1 + 11 x f2 spurs (assuming you have tabulated from -11, ..,
0,, .+11 multiples both horizontally and vertically) this would be the 22nd order spur and buried deep in thermal noise.By varying f1 and/or f2 you might even find slightly lower values for f1 and f2 with acceptable spur distribution.
yes, COUNTING on that!
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