I think I have enough of the design there to get the idea across.

I am trying to make a trans-impedance amplifier that has basically zero phase shift at 100KHz to 200KHz. The JFET in the front end is needed to get the noise low enough. Can anyone improve on this topology?

R1 is forced to be largish by the need to keep the noise current well below 1pA/sqrt(Hz).

Composite TIAs never work as well as one expects. For high feedback resistor values, I'd far prefer something like a BF862 follower running into an LT1028, with the drain bootstrapped and a sub-Poissonian current source in the source lead. That'll get rid of the input capacitance, which will help noise and stability a lot. (Depending on your accuracy requirement, a BFT25A might be a better input stage.)

You might very well need a common-base stage in the front, in order to keep the Cdiode*Rf phase shift from being a problem--feedback reduces it, but it doesn't go away altogether.

Once your sensitivity to all the temperature-variable parameters (e.g. GBW and Cdiode) is reduced, you'll know the phase shift vs frequency quite accurately, so a little lead-lag network somewhere will trim the last couple of degrees of phase nicely, I expect, without needing trimpots or other such things.

100KHz to 200KHz puts us right in the band where the LT1028's input noise rises up to about 3nT / sqrt(Hz). Wouldn't I be better off with an LT6201?

The gm of the BF862 is less than 1E-3 so the input noise current of the LT6201 would push the noise up to 2.2nV / sqrt(Hz) or did I miss something?

I assume you are suggesting a non-inverting first "op-amp" made from a source follower and an op-amp. The second stage can then be the inverting op-amp as I have it now.

True--I should have remembered. I've been designing with the ADA4898-1 recently, which is a lot like the LT1028 except with 45 dB lower DC gain, no guaranteed specs, and no funky noise peak AFAIK. It's also a bit slower and much cheaper.

The gm of a BF862 is 0.035 minimum, so the a few picoamps of noise current aren't going to be noticeable. Anyway, I thought you said you needed a large FB resistor--1 pA/sqrt(Hz) is 16k or thereabouts, which has a 1-Hz noise voltage of over 16 nV, so even 2 nV is a nit--about

0.07 dB. Of course it does cost you quite a bit of supply current, but that's life at 300 kelvin.

I really don't like using two op amps inside the same FB loop. It's usually done in an attempt to wring the last 0.5 dB out of some performance metric, and IME it always trades off something else that winds up dominating. The strange open-loop transfer function usually messes up the settling time something awful, for instance, and it's also common for input nonlinearities to be much worse than expected, especially in the presence of overshoot, which photodiode amps are prone to on account of the diode capacitance.

What sort of photocurrents are you looking at? Below 10 uA?

My main suggestion is to overdesign the first stage so you know exactly what the phase is doing between 100 and 200 kHz, and then put in a little (fixed) RC tweak to flatten it out. It should be only a few degrees.

Only if you use the wrong circuit topology. A common-base stage gets rid of that nasty noise peak very handily. What's your PD capacitance?

Understood. If you tell me a bit more about your photocurrent and photodiode choices, I can be of more help. Have you thought about the _optical_ error sources? Generally speaking, measuring an optical signal to an absolute accuracy of 1% is good going.

The part of the signal I care about is only one frequency. There are other frequencies there but they are all harmonics and I would really prefer it if they went away.

My transfer function has very little peaking. It rises by about 3dB starting at about 5MHz and then starts down without too much drama.

The photo current will be somewhat over 10uA but certainly under

100uA. 30uA is my current estimate.

rees.

So far I've got a few degrees. I need to make two that match to better than 0.1 degrees. I think this means that the 10 turn pot will appear somewhere in the design to balance them.

Hmm. With a 50k feedback resistance, and assuming you mean the overall closed-loop voltage gain, that comes out to a total summing junction capacitance (amp, pads, and photodiode combined) of something like 0.6 pF, which is a bit on the small side. Is that perhaps the transimpedance or the open-loop transfer function of just the amplifiers that you're quoting?

50k is overkill for that photocurrent--I suspect that you might be making life unnecessarily difficult for yourself. At 30 uA, 6k will get you within about 1 dB of the shot noise. That'll also push the noise peak 8 times further out in frequency, which is a big help.

If a fixed tweak isn't good enough, you can probably do just as well by controlling the photodiode reverse bias with a DAC driving a capacitance multiplier. That'll change the phase shift without causing a lot of noise. NB: the PD bias supply's noise looks just like TIA voltage noise, and gets multiplied the same way.

Have you thought about the phase shifts due to the photodiode RC?

"NB: the PD bias supply's noise looks just like TIA voltage noise, and gets multiplied the same way. "

Phil, Is there a simple way you can explain this to me. (I'm going to have to go and measure it.) I always thought that the bias voltage simply reduced the capacitance.

If the diode has a capacitance Cd, the TIA's voltage noise e_N jiggles the summing junction end of the PD, which causes a 1-Hz noise current of e_N/(2*pi*f*Cd) to flow into the summing junction.

If the bias supply has voltage noise, it jiggles the other end of the PD and causes noise current to flow in just the same way, in just the same amount.

Reverse bias helps a great deal, but it has to be _quiet_ reverse bias.

I haven't followed the whole thread, so I don't know whether it will fit, but I've used with good success active feedback in a somewhat similar case (needed stable sub milli degree 30dB amplifier @ 100kHz).

It will cost you a sqrt(2) noise voltage degradation though.

Thanks Phil, I realized how 'stupid' this question was this morning. (In other words I figured it out for myself.)

Isn't keeping the phase shift really small the same as keeping the gain really flat? (Kramer's-Kronig relations and all that.) And you'll need to keep the gain flat way past 200kHz to keep the phase shift small at 200kHz. Is it a Bessel filter (Q) that has minimum phase shifts?

For a minimum-phase network, yes it is. Lumped element ladder networks are minimum phase, but anything with actual time delay isn't, and neither are all-pass networks, which have phase delay but no amplitude rolloff.

To have a small phase shift, you need more than just a flat pass band. With a lot of poles, you can make a flat passband but have a significant phase shift within the band.

Thanks, I guess I was thinking about circuits that had some gain (or loss?) in them. I'm not very clever and so I would take a brute force approach. For flat gain and no phase shift keep all the poles far away from the band of interest.

OK I guess I can imagine that. (the all-pass filter example from above.) So is flat gain a necessary but not sufficient condition for small phase shifts. Or to ask it another way, can you build a circuit that has gain changes, but no phase shifts? (Not that I can imagine a use for such a thing.)

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