If you want the "Rolls-Royce" of an analog sin-generator fully modulable and with instantaneous peak amplitude detection which will even work at lowest frequencies without transient behaviour, I recommend a state-variable design with vectormeter to determine the peak amplitude. The design incorporates 4 multipliers and 6 opamps. I am too lazy to draw it up tho.
-- ciao Ban Apricale, Italy
Take a look at the oscillator in the AD633 data sheet. It might not be low distortion, but its a good start.
Steve Roberts
I did it in parts: .---------------------------------------------. | | | ___ C|| ___ C|| ___ ___ | out o-|___|-o---||-o-|___|-o---||-o-|___|-o-|___|-o--o | R | || | R | || | R1 | R1 | | | |\\ | | |\\ | | |\\ | | '-|-\\ | .---o-|-\\ | '-|-\\ | | | >-o | | >-o | >--o | .-|+/ | | .-|+/ | .-|+/ | | | |/ | .-. | |/ | | |/ | | === | | | === | === | | GND | | | GND | GND | | | '-' .-. | | | '-------->| |pot | | | 10R | | | | o '-' | '--o cos | | sin '---------------'
Here is the principle. The differential equation of an oscillator d2/dt^2*Vout + 2*gamma d/dt*Vout + omega^2*Vout = 0 has the solution Vout(t)= Vout_peak* e^(-gamma*t) * sin(sqrt(omega^2 - gamma^2)*t) with gamma=0 (pot in the middle) we have an sin-oscillator. with gamma=positive we have a rising amplitude and with negative a dampened one.
Since it's easier to integrate with opamps we integrate twice: Vout + 2*gamma S Vout dt + omega2 SS Vout dt^2 (S=integral) Thats what is implemented above.
Now we can compute the Vout_peak value: Vout_peak = Vp= sqrt(sin^2 + cos^2) by arranging the formula a bit this can be done with a single multiplier: Vp = sin^2/(Vp + cos) + cos we get a DC voltage from which we can subtract the reference and replace the pot with another multiplier.
To vary the frequency we can replace the two R with a dual pot or with two more multipliers.
-- ciao Ban Apricale, Italy
For slower slewing applications, what is your criticism of using a pair of inverters to allow simple full wave rectification of the sin and cos outputs, and either average the sum of those or peak follow and average as the amplitude feedback. If the frequency changes only over many cycles, a pair of LED driven photo resistors may match well enough to replace the other two multipliers, also, and let the AGC compensate for the mismatch.
No criticism, except that low THD is difficult without low ripple on the rectified dc. A.D Ryder in a Wireless World article in Jan 1981 did a similar thing with a 3-phase oscillator with fw rectification. His first stage had a THD of 0.42%, and the downstream stages reduced that to below 0.1%. The OP wants 0.01%.
In July 1982 there was another similar article from a group in Canada where they used a 2-phase oscillator and resistively generated 8 phases for rectification. A large mess of opamps, diodes, and resistors, but under certain circumstances they produced a THD of less than 0.01%.
There was a 1974 paper that apparently discussed this use of multiphase rectifiers, with calculations showing the relationship between the number of phases used and the resulting THD. I was not able to obtain a copy of that paper in the early 80's. It was:
"Fast amplitude stabilisation of an RC Oscillator." E. Vannerson and K.C Smith. IEEE Journal of solid state circuits, vol 87, SC-9 No.4, pp 176-179.
-- Tony Williams.
Thanks for the discussion, but doesn't the effect of the number of phases depend strongly on the amplitude response time, in number of output cycles? If the variable frequency needs to vary only slightly (as it does in this case and, perhaps slowly (I didn't read how fast it has to be able to change), it seems that an elliptical filter with its notch right in the middle of the range of ripple frequency, you could get a pretty low ripple gain control output.
AFAIK the intention is to get a fast-response rectification, slewing fast in both directions. This gets a stable control loop.
Multiphase rectification is one way of avoiding the smoothing capacitor that slugs the negative slew.
If some way can be found of doing a fast single phase rectification then that may allow new control methods for all single phase RC oscillators.
-- Tony Williams.
No amplitude-stabilization feedback loop?
--
Thanks,
- Win
The stabilisation pot is replaced by a multiplier, which in turn is calculating gamma x Vout. I'm gonna draw that part as well later.
-- ciao Ban Apricale, Italy
.--------------. .---| multiplier | .-----. sin | | | | add | Vout_peak o---o sin^2/(cos+Vout)------| + o----o----o | | | | add | | '---| divide in | -- --' | '--------------' | | cos | | | o---o----------)-----------------' | | | | | .--o--. | | | | | '-------| + |----------------------' | add | '-----' to 10R Vout_peak Vout o o o | | ___ ___ || | | '-|___|-o-|___|-||--. | .----------------. | R2 | || | '---| multiplier | | Vref ___ | |\\ | | | | .--|___|-o-|-\\ | | Vout x gamma |-----' | R2 | >------o-------------| | /-\\ .-|+/ gamma | | ( ) | |/ '----------------' \\+/ === | GND === GND (created by AACircuit v1.28 beta 10/06/04
-- ciao Ban Apricale, Italy
Ban didn't say there wasn't. I find this way of instantly obtaining the amplitude pretty nice since it relieves you from having an additional very LF pole in the stabilization loop, just relying on the oscillator's filter amplitude response. However I wonder how well this could behave for a very low distorsion oscillator.
-- Thanks, Fred.
That circuit is largely a wish unless you can show that any particular amp puts the 3rd harmonic 80dB down. Then the multiplier is being asked to process a 0/0 type of input which is more wishful thinking.
Fred Bartoli wrote: [sin**2 + cos**2]
Error in typical analog multipliers (MPY634, etc., the $5-$10 variety) is typically quoted at 2% full scale. When operated as squarers some of this cancels out and 0.3 (hi-grade) to 1.2% (low-grade) is what I see in the MPY634. I suspect there will be further cancellation from adding sin**2 and cos**2 since they are out of phase.
But think what happens if you have a truly constant-amplitude sine wave, and then some distortion in your squarers. The ripple due to error in the squarers will only be a percent, and it will be at twice the frequency of the oscillator under some pessimistic assumptions about the nature of the distortion, and at four or eight times the frequency under some more realistic assumptions about distortion. This is a lot better for developing AGC than a half-wave or even full-wave rectified sine wave.
Now, Ban doesn't tell us about all the time constants but I will assume that to get to sub-tenth-percent distortion that you have to average the resulting sin**2+cos**2 over a period or so (he shows a peak-measuring circuit so I'm guessing he's averaging over a couple cycles). His sin and cos oscillators are expressly designed so that as you slew the frequency, the amplitude will remain constant within the precision of the multipliers/dividers there so the gain tweaking is there only to correct for the error in those multipliers.
He doesn't explicitly say so, but I'm assuming that whenever he uses a multiplier he uses the most linear input to carry the oscillation and the least linear input to carry the gain factor. In the medium-spec analog multipliers the more linear input has a "typical" 0.01% nonlinearlity so this is the right ballpark.
Tim.
The way the vector is calculated is very similar to an RMS calculation and can probably be done with a dedicated RMS chip. It is not a 0/0 devision but a 1/1, which should be the most accurate. There is no averaging since sqrt(sin^2+cos^2) gives a DC peak value. It is just a cartesic to polar transform. The control loop can be very fast, unlike a peak value obtained by rectifiers.
Since changing the two Rs for a frequency modulation doesn't involve any energy change in the oscillator it constitudes an almost ideal fm-modulator. A cheap variable transconductance VCA like SSM2164 could be used for that, which would give even a logarithmic characteristic for the frequency control. With the remaining two VCAs a variable lowpass could eventually reduce higher harmonics at the output. It should be possible to even cover the entire audio range without changing caps, but I haven't pondered about that to be sure.
-- ciao Ban Apricale, Italy
OK, all the discussion about sin/cos oscillators and a vector voltmeter for amplitude determination was interesting, but it's not what I actually implemented. (It does come back to answer a different question I posed in the past year about starting two oscillators with a known difference in phase, though.)
Here's what I actually did. I may have cheated by actually using some elements of DDS but not actually using a numerically controlled oscillator.
VCO from an HC4046 running at 8 times the output frequency. Output frequency range was 40-80Hz so the VCO runs at 320-640 Hz.
A divide-by-eight ring counter in HC CMOS.
Ring counter decoded outputs scaled with precision resistors.
Output of ring counter run through 8 poles of VCVS Butterworth low-pass filtering with breakpoint at 100Hz. In retrospect I could've set the breakpoint higher I think (but see below about actual THD measurement).
Result looks really good on a scope. One gotcha: passband ripple in the low-pass filter shows up as variations in output amplitude at the few percent level as I sweep/step the oscillator frequency.
Has the advantage that a digital phase output is available without discriminating on the analog level. Whjen I use this with the phase comparator on the HC4046 to phase-lock to a crystal, I see phase errors in the 10's of nanoseconds, which is not too shabby for an oscillator in the low audio frequency range. To get phase noise that low I had to regulate +5V right at the HC4046. Have not tried a "low-noise" regulator, it's just a 78L05 at this point.
Of course the digital phase taken off the ring counter is not easily related to filtered (analog) phase.
I don't know much about discriminating on analog phase in a good way. The slew rate as the analog signal (say 60Hz at a few volts amplitude) goes through zero is about 100V/sec, which means 100uV/usec. I'm guessing that to get 100nsec phase on the the analog signal that I'm sensitive to noise at the 10 uV level? Is that hard? It doesn't SOUND easy.
I have NOT officially measured the distortion yet.
Tim.
It *is* a 0/0 division at the 180o points of the cos. The circuit is what is called "implicit"- another name for circular justification by assuming if the output is what we want then output is what we get.
There is a 0/0 division exactly when the output is 0-xing, so it doesn't matter really. for that short moment we could sample/hold the last exact value.
-- ciao Ban Apricale, Italy
martin
In article , Tim Shoppa wrote: [....]
Measure the output and servo the supply voltage on the shift register chip. You may need to add a CD4049 buffer in the clock signal if you swing the Vcc very far.
The servo can be very slow if you only change frequencies very slowly.
-- -- kensmith@rahul.net forging knowledge
After pondering a bit about the principle, it should be possible to just square the sin and cos and add them, without the root, since we regulate for constant amplitude. We could also rectify the cos before to avoid the division by zero. Of course the oscillator opamps and caps need to be of exeptional quality, but the Walt Jung article someone quoted makes me confident to get the required distortion figure even without post-filtering.. Of course the amplitude servo can be slowed down so there is some averaging over several periods. Now with 10*R the time to reach e-times the amplitude is 20 periods, this is only needed for very fast sweep. We could also use a 30*R for adding gamma and get the amplitude ripple more down. A similar circuit was used by Neutrik or Bruel&Kjaer back in the analog days. Fred, what do you think?
-- ciao Ban Apricale, Italy
ElectronDepot website is not affiliated with any of the manufacturers or service providers discussed here. All logos and trade names are the property of their respective owners.