# Ideas on causes of excess noise in transimpedance pre-amp?

• posted

I'm trying to understand the noise I measure in a couple of versions of a trans-impedance pre-amp that I'm using with a quartz tuning fork, such as is used in digital watches, but removed from its can and used to measure acoustic signals from a laser beam focused between the tines of the tuning fork. The output noise spectra do not match what I calculate using the Johnson noise of the feedback resistor, the specified voltage and current noise of the op-amps, the measured feedback capacitance and my estimate of the capacitance at the input of the op-amp.

The initial version of the trans-impedance amp used a Burr-Brown OPA656 with a 10 Mohm feedback resistor on a 2-sided board. I measure an output noise of 410 nV/rt-Hz at frequencies below 1 kHz, which matches the Johnson noise of the 10 Mohm feedback resistor, but the noise floor starts increasing above 1 kHz. This increasing output noise floor is qualitatively similar to that due to the voltage noise of the op-amp amplified by the noise gain. However the numbers don't workout. The specified input capacitance of the OPA656 is 4.5 pF common mode, 0.7 pF differential. There are the capacitances of the 1/2" long traces from the op-amp inputs to an SMA connector soldered to the board, the SMA connector, an SMA to BNC adapter, and a bulkhead BNC connector to which the fork is soldered. I estimate these connector capacitances total about 3 pF. Adding in the shunt capacitance of the quartz tuning fork, about 1 pF, the I estimate the total input capacitance to be 9.2 pF. I measure a 3 dB bandwidth of 106 kHz from which I conclude the stray capacitance of surface mount feedback resistor is

0.15 pF. With these capacitances and the 10M feedback resistor, I calculate that the output noise should not rise above the Johnson noise of the feedback resistor by 5% until about 100 kHz.

However I measure the noise reaching 500 nV/rt-Hz at 30 kHz. (With a quartz tuning fork connected that is still in its can, there is a noise peak at the fork resonance of 32764 Hz, but this noise peak has a width of 0.4 Hz and doesn't not contribute significantly to the noise spectrum except within about 20 Hz of the resonance frequency.)

This pre-amp has another problem, the trans-impedance gain varies with the equivalent series resistance of the tuning fork's equivalent series RLC circuit. The fork series resistance changes with pressure and temperature, so the system gain changes with gas pressure. The change in trans-impedance gain with source resistance is well fit by a model using a finite open-loop gain of 62 dB for the OPA656 which is in fair agreement with the "typical" open loop gain of 65 dB up to about 100 kHz.

So to reduce the variation in trans-impedance gain with source resistance, I decided to replace the OPA656 with the OPA657. The OPA657 has a higher typical open loop gain (75 dB) and a larger gain bandwidth product, but the same specified input capacitance and resistance as the OPA656, the same current noise (1.3 fA/rt-Hz), and slightly lower voltage noise (5 nV/rt-Hz for the OPA657 and 7 nV/rt-Hz for the OPA 656). To my surprise and disappointment, with the OPA 657, the noise increased at 30 kHz from 500 nV/rt-Hz to 600 nV/rt-Hz and peaks at 50 kHz. With the OPA656, the noise kept increasing to 100 kHz, the upper limit of my spectrum analyzer.

To further confuse me, another version of this pre-amp was made with a

4-layer board that has provision for doing active bandpass filtering. We found that the trans-impedance input stage with an OPA657 has twice the noise at 30 kHz as the previous board with an OPA657. This higher noise level is present when the components for filtering are removed. Even when a quartz tuning fork is soldered to the 4-layer circuit board, the noise level near 30 kHz is still about 3 times the Johnson noise level of the feedback resistor.

I have measured the noise spectrum at the power pins for the OPA657 and it is flat at 200 nV/rt-Hz from about 128 Hz until it starts rolling off at about 40 kHz. I have also looked at the output of these pre-amps with a 400 MHz bandwidth scope and can not find evidence of oscillation. I found a problem with current leakage on one of these boards that was fixed by scrubbing with isopropyl alcohol and cotton swabs, so we've periodically cleaned these boards without any reduction in noise since that first time.

The one other obvious difference in the two circuit boards is that on the

4-layer board, the ground plane extends under the op-amp and perhaps under the traces from the op-amp inverting input to the fork connections, while the original 2-layer board does not have a ground plane under the op-amp and the inverting input.

My basic questions are:

1) What could be the cause of the excess noise that became significantly worse when switching to a faster and higher gain op-amp that has less voltage noise and that depends on the layout of the circuit board?

2) What things should be done in the layout of the next circuit board to minimize the noise near 32 kHz for a trans-impedance pre-amp with gains of 1 to 10 Mohms?

Thanks for any suggestions, Bret Cannon

• posted

The corner at 1kHz implies a capacitance to ground of 16pF - a bit more than your estimated 9.2pF.Printed circuit tracks are microstrip transmission lines, and the capacitance per unit length depends on the width. 50R tracks are fairly wide and run at 3.8pF/inch (150pF/metre).

75R tracks look like regular trances and run at 1.7pF/inch.

Philips use 2.2pF/inch in one of their application notes on using high speed logic.

The extra 6pF suggests that you have neglected to allow for some three inches of track over ground somewhere in your system.

That sounds wrong. If the gain corner is at 1kHz, the noise gain at

30kHz is 30, so the 7nV/root Hz input noise of the op amp will look like 210nV/root Hz at the output at 30kHz, 700nV/Hz at 100kHz

The root means square sum of 210nV/root Hz and 400nV/root Hz is

452nV/root Hz.

The noise spectrum of the amplified inut noise won't be white, because the high frequency components are amplified more than the low, so rms summation may be over-optimistic.

The stray capacitances on the printed circuit board seem to be higher than you have estimated. Have you tried measuring the stray capacitance on an unpopulated printed circuit board?

Minimise the stray capacitance - shorter tracks, narrower tracks, remove ground plane under the crucial tracks, as recommended in the Burr-Brown application notes. Using a two-layer layout on the outer layers of a four layer board reduces the distance to the ground plane by a factor of three, and roughly triples the capacitance of any microstrip tracks, even before you add in any extra ground plane.

Make sure that you have both the recommended 100nf decoupling capacitor to ground close to the op amp's power pins, and a couple of uF of tantalum capacitorless than an inch (25mm) away, as recommended by the application notes. I tend to throw in a ferrite chip/bead between the capacitors and the power rail - they are fairly cheap, don't introduce any DC drop, and - being lossy non-wound devices - look resistive rather than capacitative up to very high frequencies. The of the order of 1uH of inductance can resonate with the decoupling capacitor, but as long as you have a few uF of tantalum capacitor present, the resonance is around 100kHz and the impedance at resonance low enough - at around

0R6 - that the ESR of the tantalum bead will kill any peaking.
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Bill Sloman, Nijmegen```
• posted

I measured the stray capacitance on both boards and the connectors. Both are larger than I had estimated, but not enough to explain my measured noise. The first board with connectors and the op-amp input capacitance is

16-19 pF while for the second board the capacitance is 18-21 pF. The measured capacitance on the first board now gives good agreement when the op-amp is the OPA656 (at 33 kHz the model gives about 475 nV/rt-Hz while I measure 500 nV/rt-Hz.

These larger capacitance values still leave unexplained why when I changed to the OPA657, which has the same current noise and input capacitance specs as the OPA656 and slightly lower voltage noise (5 nV/rt-Hz versus 7 nV/rt-Hz), the noise at 33 kHz increased to 600 nV/rt-Hz.

Also unexplained is why with the second board and an OPA657, the noise is

1000 nV/rt-Hz, which is more than twice what I calculate.

I did confirm that the power pins have 100 nf decoupling capacitors, 10 uF capacitors within 1/4" of the power pins and a ferrite beam between the capacitors and the power rail.

Bret Cannon

• posted

In this situation, you should measure your various opamp's actual voltage noise, rather than rely on a spec.

BTW, a possible noise source that hasn't been mentioned is a falling opamp power-supply rejection at higher frequencies, and high-frequency power-supply noise. Usually your bypass caps help reduce the latter, etc., but it's worth examining.

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Thanks,
- Win```
• posted

"Connectors" implies lengths of coaxial cable, with a capacitance of about 100pF/metre. I presume that this is included in the measured capacitance. What are you using to measure your capacitances?

The OPA656 is going to run out of gain at about a MHz, at a gain of about 100, set by the 0.15pF parallel capacitance of the 10M resistor, so the output will have some

800nV/root Hz of random noise. The closed loop bandwidth is about a MHz, so the rms noise will about 800uV, about 5mV ptp.

The OPA657 has six times the bandwidth, so the broadband noise on its output will be higher by root six, say to 12.5mV peak-to-peak.

This makes the output relatively noisy. Do you have a damping resistor on the op amp output, to isolate it from any capacitative loads? The data sheet recommends 100R for a 10pF load, progressively reducing as the capacitative load increases (see the curve at the bottom of page 7 of the OPA657 data sheet). The op amp may not oscillate without it, but the closed loop gain can develop a signficant peak before it starts rolling off, which might boost the noise to the point where the output stage starts getting non-linear and producing difference products between the noise signal and harmonics in the tuning fork output.

The extra bandwidth of the OPA657 does make it better at picking up externally generated noise.- once, early on in my career, I found that a significant part of the "noise" on a circuit I was looking at was the local AM radio transmission. Somehow the circuit had managed to configure itself as a moderately effective AM receiver. More careful shielding dealt with that.

Sounds good.

Bill Sloman, Nijmegen

• posted

For the first board, the connectors are an SMA bulkhead connector soldered to the board, an SMA to BNC adapter, and a BNC male union. To measure the capacitance, I connect a Fluke 4 1/2 digit multi-meter (I don't remember the model) via a twin banana-plug to BNC adapter. The capacitance reading of the Fluke with the banana-plug adapter in place is a quite stable 0.067 nF and when I connect the first circuit board with its connectors, the capacitance reading increases to 0.074 nF if the red terminal of the multimeter goes to the center pin of the BNC. If I reverse the banana to BNC, then this last reading is 0.077 nF, while the "zero" is still 0.067 nF. I measure the capacitance of a tuning fork soldered to an isolated BNC bulkhead as 4 pF independent of polarity.

I see noise spikes on the spectrum analyzer at 19 kHz and its multiples and at 43 kHz unless I shield the input. (The 19 kHz is from switching supplies in the lab and I think that the 43 kHz is from the fluorescent light ballasts.) However, with several layers of aluminum foil connected to the case of the pre-amp or putting a metal box around the input, these noise spikes disappear into the noise floor. I haven't used an rf spectrum analyzer to look for any pickup above 100 kHz, but I don't see anything on a

400 MHz bandwidth scope.

The output of the OPA656 and OPA657 on both boards has a 50 ohm resistor in series to the output connection and a 100 ohm resistor to ground, which I'm told reduces oscillation problems.

By the way, I'm measuring the noise with a Stanford Research FFT spectrum analyzer, so I assume that the filtering on the spectrum analyzer takes care of the noise at frequencies above 100 kHz. Is this wrong? If not, then why would the wider gain bandwidth of the OPA657 increase the noise I see with the spectrum analyzer?

thanks, Bret Cannon

• posted

It was my understanding (perhaps wrong) that to really characterize the noise of an op-amp correctly , you need 4 values....

the current noise of the + input,

the voltage noise of the + input,

the current noise of the - input

and the voltage noise of the - input.

Depending on what you have connected to the op-amp, these 4 numbers will have varying significance. If you are connecting Hi Z sources, the current noise becomes very important and the voltage noise less so.

see:

Mark

• posted

that bit me a while back. I had 10dB of *GAIN* thru the supply at 100kHz.

Cheers Terry

• posted

The 50R resistor in series with the output connection os good - not only does it isolate the amplifier from any capacitative load, but it also acts as a series termination for driving 50R transmission lines (like regular coaxial cable).

The 100R resistor to ground is a mistake - the amplifier is specified for driving a 100R load, but that is your 50R source (series) termination resistor, plus the 50R parallel terminating resistor at the other end of your 50R coaxial cable. Adding a 100R resistor to ground in parallel with the cable load doesn't do anything useful, and overlaods your output stage.

Whoever used it to kill and oscillation didn't know what they were doing - the extra load may kill the amplifier's gain enough to stabilise the loop, but at the expene of crippling the amplifier.

Get rid of it.It may be indirectly contributing to your noise problems.

The wider gain bandwidth of the OPA657 is going to mean that there is more wide-band noise on its output than there is on the output of the OPA656. This means more wide-band current flowing into and out of the power supply pins, and more opportunity for high -frequency noise components to mix and generate low frequency difference products that may add to your Johnson noise sources.

If you are over-loading the output stage, you may be forcing that to operate in a voltage current region where it isn't all that lnear, and the non-linearity can allow high-frequency components to mix and generate extra low frequency noise..

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Bill Sloman, Nijmegen```
• posted

I'm pretty sure the 19kHz you see is coming from the radiated magnetic field from the SRS analyzer's own CRT sweep circuits. But there's very little energy in these narrow spikes and they can be safely ignored.

Did you see my point about measuring the opamp's actual voltage noise density, rather than relying on a datasheet typical spec? That's an important step, as is checking the opamp's power-supply high frequency rejection, and your power-supply's noise spectrum.

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Thanks,
- Win```
• posted

Thanks for the response.

I measured the input voltage noise density of the OPA657 on the first board today and it matches the datasheet value of 4.8 nV/rt-Hz at 100 kHz and was only slightly above 5 nV/rt-Hz at 10 kHz.

I've also measured the power supply noise densities at both the positive and negative and the are well below the output noise density from at least 100 Hz to 100 kHz. There is a slight noise peak at about 60 kHz on in negative input power to the metal case, but the -5 V regulator and the filtering on the circuit board seem to eliminate that noise bump.

Bret Cannon

• posted

The current noise spec is 1.3 fA/rt-Hz and this is much less than the Johnson current noise of the 10 M feedback resistor of about 40 fA/rt-Hz. With the non-inverting input shorted to ground, the current noise at this input would not contribute anyway.

I have seen some op-amps with different noise specs for the inverting and non-inverting input, but the datasheet for the OPA656 and OPA657 don't mention any difference for these parts.

Bret Cannon

• posted

We removed the 100 ohm resistor to ground today and the noise spectrum didn't change.

I borrowed an rf spectrum analyzer so I could look at frequencies beyond 100 kHz and didn't see any noise peaks or oscillations up to 1 GHz. I did find that from about 200 kHz to a couple of MHz the noise is flat and matches 5 nV/rt-Hz times a noise gain of 7 pF/0.15 pF with nothing connected to the SMA connector soldered to the board. So it seems that I'm dealing with a noise peak at about 60 kHz in addition to the Johnson noise of the feedback resistor and the noise gain times the input voltage noise.

Bret Cannon

• posted

I saw your post and replied to it. The voltage noise density that I measured is an excellent match to the data sheet in the range of the SRS analyzer.

The noise on the power supply pins of the OPA657 on the board is about 35 nV/rt-hz at 60 kHz where the noise peak is, and is falling. Do you think I need to check the power supply noise spectrum at higher frequencies than the

100 kHz upper limit of the SRS analyzer?

Perhaps I misunderstood your point about checking the op-amp's power supply high frequency rejection. Do you think that I should measure these values on the op-amp on the board? Is there an easy way to measure this without pulling out all the power supply filtering on the board?

Thanks, Bret Cannon

• posted

Well, by process of elimination is seems we're pointed back to the circuit's capacitance. I don't see an explicit mention of this noise source in your posts, except for "the model," which you didn't specify. The noise we're talking about is what I call "e_n-Cin" noise, an apparent noise current i_n = e_n Cin 2pi f, where e_n is the effective opamp voltage noise density, including PSRR stuff, etc., and Cin is the sum of all the capacitances at the summing junction node, including the sensor, all the internal input-stage capacitances of the opamp, etc. Note this noise source rises with frequency, like the one you observed. The e_n-Cin noise current is multiplied by the amplifier's trans-resistance and is seen as a rising voltage-noise density on the output. The latter assumes a flat transimpedance, of course, which may be incorrect and has to be separately tested with calibrated input current signals. I say calibrated, because you cannot necessarily simply create small test currents with a resistor, due to the resistor's parallel self capacitance, which is generally about 0.2pF or so.

We're looking for any peaking of the amplifier's transimpedance, which will occur if the oamp's f_T is insufficient or the feedback resistor has not been properly bypassed, f_T > fc^2 2pi Rf Cin, where fc is 1/2pi Rf Cf. For Cin = 20pF+5pF (opamp) and Rf = 10M and an fc of 60kHz we see f_T must be greater than 5MHz. Checking on the OPA657 we see its spec is 1600MHz. OK, that doesn't look like a problem. :-) BTW, at 60kHz the 10M feedback resistor and its strays must have much less than 0.26pF of capacitance to still look like a resistor and not a capacitor. What have you done about that? Did you tell us what the overall frequency response of your transresistance amplifier is supposed to be? Have you measured it? How, exactly?

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Thanks,
- Win```
• posted

The noise model is the "e_n_Cin" noise added as the square root of the sum of the squares to the Johnson noise of the feedback resistor. The Johnson noise is rolled off by the stray capacitance of the feedback. In addition, I've included the finite open loop gain as a function of frequency of the OPA657 rather than taking the limiting case of infinite gain.

I measured the frequency response using the SRS analyzer, but with using an old switched attenuator box to reduce the drive voltage to about 1 mV. This drive voltage goes to channel 1 of the SRS analyzer and drives a thru-hole resistor soldered into a Pomona box. I've used resistance values from 1 M down to 25k, though most of the time I've used either a 91k or 37.4k resistor, since this matches the series resistance of the quartz tuning forks at the pressure of interest. The transimpedance gain with both 91k and 37.4 k is down about 2.5 dB at 100 kHz. I calculated the 3 dB frequency is about 106 kHz, from which I calculate the capacitance of the feedback as 0.15 pF. The frequency response shows a little peaking, less than 1 dB, before it starts to roll off.

BTW, I measure the transimpedance gain varying with source resistance, which is very well fit with a model using a finite open loop gain for the op-amp. The parameters for a best fit to my data are an open loop gain of 72 dB versus 75 dB from the datasheet and the model feedback resistance matches the actual feedback resistance to 1%.

Also, yesterday I borrowed an rf spectrum analyzer to look at the noise above 100 kHz. I didn't see any noise bumps or oscillation up to 1 GHz. I did see a flat noise density from about 200 kHz to a few MHz that matches (5 nV/rt-Hz) x (7 pF/0.15 pF). It seems to me that what I'm fighting is a noise peak at about 60 kHz on top of the Johnson noise and the "e_n_Cin" noise.

Did you get my last post from last night asking about measuring PSRR on the board? Terry Givens' posting of having seen 10 dB of gain at 100 kHz in an op-amp has me thinking that 35 nV/rt-Hz noise on the power pins at 60 kHz may not be enough to eliminate that noise source.

Thanks, Bret Cannon

• posted

I'm too buried to go through all the posts on this topic, so apologies if this has been suggested before....but a 60 kHz peak can be caused by a linear voltage regulator with the wrong bypass cap. The output of a

7805 looks inductive, and the bigger the bypass, the bigger (and lower in frequency) the noise peak is.

You might try putting a 5-ohm resistor in series with the regulator output and see if that makes life better.

Cheers,

Phil Hobbs

• posted

Come on Win - pay attention. In his original post Bret had already assigned a value of 0.15pF to the parallel capacitance of his 10M resistor, based on the frequency dependence of the closed loop gain of the op amp.

Bret does seem to know his physics, though he isn't particularly sophiistcated about electronics. At the moment he seems to be happy with the noise figures for the circuit built around an OPA656, but the nominally quieter OPA657 produces more noise than the OPA656, and he can't find an explanation..

The only explanation that I've been able to come up with involves handwaving about the difference between the 5mV peak-to-peak broad-band noise he ought to be seeing at the output of the OPA656, and the 15mV ptp I'd expect at ithe output of an OPS657.

The fact that the removal of the 100R to ground at the output didn't make any difference to the noise level suggests that this explanation doesn't work.

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Bill Sloman, Nijmegen```
• posted

it was a TL064, 1MHz GBW from ST, very cheap and very low power. I think your opamp will be a bit better :)

Cheers Terry

• posted

Just to clarify this "flat noise" bit. Once you're above the e_n-Cin noise threshold, the noise density goes up with frequency. But above 100kHz you're also above the transimpedance rolloff frequency, where the gain is dropping with frequency, thanks to the feedback resistor's self capacitance. These two cancel and you see a flat output spectrum.

• But, this assumes the opamp has plenty of extra bandwidth, per the formula I posted, so it doesn't also contribute to the rolloff. For folks who have chosen a barely adequate opamp GBW, the rolloff will be -12dB/octave rather than -6dB, or perhaps even more. Generally this sharp rollof is a good thing. Certainly it helps to reduce the wideband rms noise.
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Thanks,
- Win```

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