Core power handling Capability

It is from this application note from On Semi.

This is pretty well all they have for the transformer.

"Transformer

Below are the key parameters you will pass to your transformer manufacturer to help him select the right winding size and tailor the internal gap: Maximum peak primary current, including 160 ns propagation delay: 1 / 0.33 + 374 × 160 n / 700 m = 3.2 A Maximum primary RMS current at low line: 1.6 A Maximum secondary RMS current: 6.9 A Primary inductance: 700 mH Turn-ratio, power section: Np:Ns = 1:0.166 Turn-ratio, auxiliary section: Np:Naux = 1:0.15"

I'm not exceeding any of those ratings upto 90W with a safety margin. If I had to guess and I do it may be either a ETD34 its about 1/2 a cm longer then an ETD29 core that I have or an EE32.

Yes your probably right. I will just hook it up and monitor for saturation and temp rise.

Components have improved since they did the AN I'm using a FET with

The spread-sheet assumes that you are reconfiguring turns, gap and wire guage for the different operating frequency.

Simply increasing the frequency may reduce flux peaks, but core loss will not reduce proportionally if the frequency is increased at the same time. The net result is increased total loss even without an increased current through-put.

Copper losses increase with the square of the current density.

What actually happens depends on how the original transformer was designed. If you don't know the core shape, material, turns, wire gauge, voltage ratios and frequency, there's little point in speculating.

It'd be faster just to fire up the circuit and measure deltaT as the frequency-setting component is altered - if the circuit is pre-existing and you nkow no more about it.

If the circuit is not pre-existing, you'll get better results working with fewer unknowns. Transformer loss is not the only consideration as a flyback circuit's throughput power increases.

RL

The loss in a ferrite core should be mostly hysteresis loss, which is roughly proportional to frequency and square of magnetic field intensity. Power throughput, at least in an oversimplified case, is proportional to the squares of frequency and magnetic field intensity.

Ideally, ratio of throughput to core loss is proportional to frequency.

Meanwhile, there are other issues:

1. Resistance of the copper will be higher at the higher frequency, approaching proportional to the square root of frequency once the "skin depth" gets much smaller than the wire radius.

1. The transformer may have enough inter-layer capacitance to cause a significant lowpass filter effect. Do you know that it will work at the higher frequency?

Or are you planning to rewind it?

1. If you rewind it and use fewer turns of thicker wire, keep in mind that the wire's resistance at the frequency in question may be closer to inverse proportional to the wire's circumference than to its cross section area due to the skin effect.

1. The switching transistor's switching loss, as a percentage of power throughput, is proportional to frequency. The transistor's rise and fall times will probably be in the tens of nanoseconds, possibly more.

Consider the energy dissipated assuming rise time times half the current initially conducted by the transistor (possibly zero) times the input voltage, plus the fall time times half the current being conducted by the transistor shortly before switch-off times the voltage that the transistor experiences during switch-off (always more than the input voltage, usually by a factor of more than 2, in flyback circuits).

Keep in mind that rise and fall times in transistor datasheets are at junction temperature of 25 degrees C and with ideal or very good driving of the transistor. In actual applications, these times are usually slower (longer periods of time).

Multiply the switching losses by frequency, add conduction losses (increases with temperature if the tyransistor is a power MOSFET), and determine if that is going to be too much heat for the transistor to dissipate. If the heatsinking is currently minimal, then it is *probably* easy to hack additional or greater heatsinking onto the switching transistor.

- Don Klipstein ( snipped-for-privacy@misty.com)

Quite frankly, I've never seen a power transformer described in this manner, nor is the list actually complete without it's surrounding article, in which topology, operating frequency, input voltage range, and output voltage are actually mentioned. It has to be assumed that the part must meet the requirements of some coordinated safety standard ~ 60950.

The article itself seems a little quirky - focussing more on control chip pecadiloes than power train. As the part is being made available for this application, there is little emphasis on practical transformer design issues.

One example; the choice of primary inductance is made arbitrarily on the basis of full load transition from complete to incomplete energy transfer mode (an irrelevant feature) at an arbitrary input voltage, somewhere (the author hopes) the psu will never actually have to run. Then this careful calculation is (equally arbitrarily) approximately doubled.

Another interesting issue is the fact that, in the end, the author somehow achieves the design spec output power only at a higher output voltage than intended. Whether this indicates that the final iteration was incapable of the design spec current and voltage, or that the author was just to lazy to complete an accurate set of drawings for his article - is a mystery to me.

You'd be better off haunting the old Unitrode seminar app notes, if you're interested in flyback transformer design iteration.

RL

Yes it is from Coilcraft; here;

Its the Z9260-AL

I've noticed that some game consoles have the exact same specs PSU wise. 16.5V, 4.2A. I never took one apart but I'm guessing they are the PSU from the application note.

Well it is written by C.Basso for Onsemi so I view like I view most application notes like commercials.

This I understand why they would do it. I don't think you would want a Flyback much above 30W operating in DCM at low line peak current could start to get high.

The reason they give for adjusting the inductance is to reduce the peak current; the controller enters a burst/ skip mode at one third full load if the peak current is high the supply would generate excessive noise. The burst mode is in the audible frequency range.

They probably tested and determined that 700uH is when the supply is tolerable noise wise.

They could have just as easily decreased the level at which the controller enters burst mode.

You noticed that too:-) I was curious about why they did that as well. They mentioned earlier in the AN about having difficulty in maintaining sufficient voltage level on the AUX winding during burst mode this might have something to do with it.

They are trying to highlight the amazing low standby power the chip can achieve. This can only be obtained with the DSS inactive and the controller being supplied from the aux winding.

I'm hoping to have the time to get a board done up some time through the week.

I'm using the NCP1217 100kHz version. I found an appnote using that controller and the same CoilCraft transformer. The power output is 85W but again it's higher output voltage at 3.5A.