Norton noiseless feedback amp calculation

I was not looking for a discussion to the n'th degree and defined down to pin point accuracy. That thumbsuck is fine and a whole lot better than nothing.

Some pointers please, a few words to put me on the right track and save hours of messing about by being able to ball park the behaviour.

Thanks Peter

From what I recall of the original Norton article, the main thing is noiseless input impedance termination in addition to gain setting. The circuit calculation is straightforward. Break it down into the hybrid-pi corresponding to your operating point- that would be the operating point computed from the equivalent DC circuit. What is your question anyway?

Write out the equations and solve them. Routine feedback.

OK. Stop complaining.

Look up Dallas Lankford's articles (Microwave Journal, May 1975) and patents (3,426,298; 3,624,536; and 3,891,934) on common-base transformer-feedback (CBTF) amplifiers. Good for low-Z inputs.

There's also been some good discussion here on s.e.d. And check Lepaisant in RSI 63-3 (1992). "Low-noise preamplifier with input and feedback transformers for low source resistance sensors" by J. Lepaisant, M. Lam Chok Sing, and D. Bloyet

If I could I would have and not wasted your valuable time and sarcastic advice.

I have the patents and these deal with a common base amplifier and feedback via the transformer. Sorry I just don't have the knowledge to write out new equations where the transformer is terminated in the emitter impedance and combine the feedback.

While Re maybe 25/Ima this must be combined with the feedback.

How?

or Zi in = Zout but my use required a high input impedance. The feed point of those amplifiers ia also the junction of the emitter winding and Rs which I have decoupled.

The above deals with the case where Rout is determined by Rin or vice versa. m*m/(m+n+1)Rs = Rl

The decoupled Rs common emitter is determined by other factors such as Re but that is not all.

I would also be interested to see what then net effect of additional degenerative feedback is on the comon emmitter circuit.

Sorry this is Africa..... If it is not available on the net I am not going to get it in a hurry or without traveling some distance and begging to use a U library.

snip

Correct but the original is a common base with signal injection at the emitter winding and external resister junction. Rin determines Rout.

I have that point decoupled and the signal is injected at the base or source as the case may be.

The use I wanted was for a SW receiver RF amplifier replacement. The original design dictates how this will be done. The input is from the preselector tuned circuit so this can't be loaded or Q will be reduced. A dual gate FET was originally used with Rd = 560 ohms. The output must match a filter Zin = 560 ohms. Freq 0.5MHz..30MHz.

It is also such a useful building block that I can think of at least a few other uses even as an output buffer because the transistor version looks like low output impedances are quite easy to achieve ie 50 ohms with reasonable gain.

I have two old valve signal generators that desperately need a 50 ohm output :-)

It's not for production because I can see that Zo will vary with the operating point and that would need to be stabilised at the very least.

My apology not all are blessed with a higher education and I am totally self taught. Hybrid Pi is Greek to me ;-0

Regards Peter

Win's been very helpful to you and I can't see any sarcasm in his response - look it up in a dictionary. He was probably niggled by your apparent inability to search for relevant stuff for yourself.

Leon

If I had not done any research and I knew how to model the circuit I would not have asked. That much is obvious to anyone who reads my original post. Win made the incorrect assumption I was being lazy and got the correct response for it. He also incorrectly assumed I had the skills to remodel the circuit and I have already stated I did not.

Possibly I was niggled by his inability to read and understand, for that I apologise. His references that I can obtain all point to the common base version because I have them already.

The only reference I can remember to the common emitter is that of Ulrich Rhode 1980s Ham Radio article and I don't have a copy and have no source of obtaining it.

I would appreciate some help, not be told to do what I have done or have stated I can't do. All the references I can find deal with the common base version as detailed in my reply. An available reference to the common emitter version would be nice or some help in remodelling.

I appreciate that the patent papers show a specific application and show the math but that is to my mind significantly different (Rin = Rout is not device and operating point determined) to what I wanted. If I knew how to rearrange it or change it I would have done so and not wasted everyone's time. Originally clearly stated.

I have monitored this group for some 20 years and have learnt much from that. I don't ask questions without at least making a large attempt and effort to answer it for myself. All I wanted was a bit of help from those far more knowledgeable than myself.

Regards Peter

Well, that what the rest of us have to do. It always amazes me how some folks who are asking for our help, and want us to spend our time giving it, can be so demanding, with details of what they are or are not willing to do on their part.

Anyway, I've added the article, Lepaisant_preamp-xfmrs_RSI.pdf on my wesbsite, at

As for the technique, it's simply substituting a transformer, with its noiseless turns ratio, for resistors in a feedback circuit. You can use the technique for all kinds of circuits, but its best value as a low-noise technique comes when you're working with very low impedances. That's because if resistors are used, they must have absurdly-low values. For example, if you're making a 100 pV/root-Hz amplifier, you may need to use feedback and bias-setting resistors, etc, with values that are under 0.6 ohms, to keep their Johnson noise from ruining your amplifier. On the other hand, if you use a transformer, you just have to keep the winding resistance well under 0.6 ohms.

OTOH, if you're making a high-impedance amplifier, then it's likely you won't need the transformer's low noise-resistance.

Now I remember, check out the s.e.d. discussions about Jeroen Belleman's amplifier designs, and his web page,

The feeback is Vo/N The sum is Vi-Vo/N

-Vo is Hfe(Vi-Vo/N) Vo/Vi = Hfe/(1+Hfe/N)

So for N = 10:1 and Hfe = 60 Vo/Vi = 8.57

Not quite applicable as can be seen by the difference in the circuit shown

Zo above is determined almost completely by the impedence seen by the feedback winding.

Correct this is essentially the same but no treatment is given for determining Zo.

Without feed back a signal injected at the feedback point would see Re/10mA Thus Zo is reduced from N*N*Re/Ic(mA) by the feedback Or for the figures I gave Zo < 100*2.5 = 2K5 ohms

Intuition says its closer to 50..60 ohms.

I can't figure how to calculate it.

Regards and thanks

Peter

Re/10mA, what's that? Do you mean V_T/10mA?

If you were using a BJT, where re = kT/qIc = 2.5 ohms at 10mA, there are several ways to show Zo = N^2 higher, or 250 ohms. I measured 150 ohms in a spice analysis, using a bc547. For a FET, use 1/gm at the operating current. Zo will likely be 5 to 20x higher, or as high as 2.5k, etc. You may want to add an emitter follower.

Explain your intuition.

You're on the right track if you use 1/gm, rather than hfe in your thinking. hfe has little effect other than Zin.

A month has gone by, why not make one and measure it?

BTW, did you get all the articles, patents, and web-links that I referenced, and even posted for you on my website last June 3rd in this thread? Any thanks due? You think?

Eek that should be 250 ohms. My apology.

Sorry I see I made several typo mistakes, should just be Re derived from the 10mA IC ie 2.5 ohms.

For Zo = 50 ohm with a follower (BJT) implies a IC of 1 mA or less. Is that a good idea with an amplifier that needs an IP3 of +20..+30dBm?

However in this application I required a Zo = 250 ohms and gain in the region of 20dB from a FET.

It is possible to tap the output winding at any integer value between 1 and N. Zo of 450 and 200 could also just as easily be handled by a simple small additional transmission line transformer on a ferrite bead. In most cases Ic can be varied by a factor 2 ie 7 to 14mA to acheive a specific Zo.

My original application dictated a minimun of components which had to be retro fitted to an existing dual gate mosfet RF amplifier feeding a 560 Ohm Zin filter. Zin had to be high so as not to degrade a resonant tuned circuit.

But it is an interesting circuit that I wanted to know more about, for knowledge and who knows.

If Zo can be adjusted by IC then small changes are possible for a one off. For production it would be a nightmare with unselected FETS.

Low Zin (BJT) can be set with C-B voltage feedback resister. Here Zin would be Rfb/N.

A test of BFR91A @ 11.4mA and N = 38:3 Emitter resister 56 decoupled by

Gut feel ;-)

OK but how does the feedback influence that? I dont know if I have a mental block on this but no amount of looking at anything has turned the light on. All I really want is a better understanding and throwing high level papers at me will not give that. Consider giving a caveman a match patent is not going to help him start a fire.

I did.

Measure it?

I wanted a ballpark method to cut down on the iterations of winding small ferrite beads and out of general interest. I tried a variable resistor (cermit trimmer) across the output winding, isolated by a capacitor 0,1uF and adjusted until I had half the unloaded voltage swing (600mV) out. Then measured the resistance. Was this what you had in mind? Vector impedance measuring equiment does not just ly around and the last I had access to was 20 plus years ago.

For the FET I got 560 ohms or close enough. Device was a BF966 @ 7mA and N was 35:3 after several iterations and a check for low frequency response. The bead used for T1 was an unknown gathered somewhere in a past life. Initial tests used a Mullard FX1115 bead. Most wound with twisted bifilar or trifilar 0.198mm wire.

I got caught by the source resister decoupling and not realising the very low impedences to be expected. That wasted some time and effort in thinking it was the transformer material and lack of inductance. The FX1115 would have done fine.

Installing in the receiver (FRG7) also produced an oscillation due to poor supply line decoupling and layout.

"Low-noise preamplifier with input and feedback transformers for low source resistance sensors" by J. Lepaisant, M. Lam Chok Sing, and D. Bloyet was available on subscription only, thanks for a copy but once again the input is via the feedback winding in common base configuration so Zo depends on Zin. This is not the case for the common emitter version with emitter feedback as shown above, please look again. It may seem obvious to you how to derive Zo from common base for the common emmitter version but it is not to me. You can point me in the direction of 1000 papers on the common base version and it will not help one little bit. If it's to much trouble to state what one has said was trivial just use a hybrid pi model, then just say so. I did try and it is beyond a gap I can't seem to cross.

I had all the patents and what was available on the net which deal with the common base version only. Thanks for the pointer to the SED discussion and the paper you so kindly made available. Ulrich Rhode gives some detail on the three configurations in push pull but not any way of determining Zo or Zin. Yes I found my copy of his Ham Radio article just discussing the configurations and possible applications after much digging around in 20 year old notes. Wes Haywood also had some stuff published in ARRL but once again this is based on the patents and common base version.

I wonder how insensitive you would feel knowing my father had passed away. Thanks for your help, effort and time it is much appreciated.

Peter

It's not a good idea to use the output impedance of an emitter follower as a matching source impedance for a filter, etc., because the output impedance is a small-signal value that can be highly nonlinear with signal swing and transistor current. Instead establish a nice low Ro value, with 2 to 4mA, etc., and then add a series resistor to drive your filter.

Yes, that's reasonable. Add the EF, you'll be happy, then the exact Zo of the amplifier stage won't matter.

I'm sorry to hear of your father. It took me several years to get over mine dying, and I was surprised to discover later that I had as a result made a direct change just then in some of the things I had previously enjoyed. I also had unexpected health responses. It's important to try to maintain a good attitude.

Not something I would usually do. I was trying to establish a Zo of 560 ohms in the drain circuit via the feeback winding in the source.

That is another typo above. Sorry.

I was in fact trying to avoid the resistive loss which would have to be added to the filter insertion loss. Since it is the RF front end, noise figure was also important as well as large signal handling capability over

N turns are the drain winding but Zo must be determined to a large degree by the source winding which sees via feedback the transformed source impedance.

I wanted a insight to the calculation of the drain Zo with source feedback transformer so I could see how well established it was and what would have the greatest influence on it.

I still have no idea how well established Zo = 560 ohms was for small and large signals. The original circuit used a 560 ohm drain resister and no feedback. Unfortunately I don't have equipment to check the filter response with both amplifier circuits or I would have done this. The possible influence of the filter reflected impedance changes on the amplifier drain are also an unknown and my thinking was feedback may improve this.

What does EF refer to? Sorry I am not with you on this.

Pops was 91 and has a good innings. He died in his sleep at my home where he has been staying for the last 5 years since my mother died. Gave up driving at 86 when he came to stay with me.

Thanks for your kind words and help.

Peter

Not something I would usually do. I was trying to establish a Zo of 560 ohms in the drain circuit via the feeback winding in the source.

That is another typo above. Sorry.

I was in fact trying to avoid the resistive loss which would have to be added to the filter insertion loss. Since it is the RF front end, noise figure was also important as well as large signal handling capability over

N turns are the drain winding but Zo must be determined to a large degree by the source winding which sees via feedback the transformed source impedance.

I wanted a insight to the calculation of the drain Zo with source feedback transformer so I could see how well established it was and what would have the greatest influence on it.

I still have no idea how well established Zo = 560 ohms was for small and large signals. The original circuit used a 560 ohm drain resister and no feedback. Unfortunately I don't have equipment to check the filter response with both amplifier circuits or I would have done this. The possible influence of the filter reflected impedance changes on the amplifier drain are also an unknown and my thinking was feedback may improve this.

What does EF refer to? Sorry I am not with you on this.

Pops was 91 and has a good innings. He died in his sleep at my home where he has been staying for the last 5 years since my mother died. Gave up driving at 86 when he came to stay with me.

Thanks for your kind words and help.

Peter

Understood. But not the best idea, IMHO.

It's certainly laudable to be careful with every dB of gain while the signal is still weak, or the signal impedance is high, etc., but once you've gone through a healthy gain stage, e.g. 20dB, you no longer need to save every dB (or even every 6dB) and other considerations take precedence, such as low distortion, accurate filter properties, etc. By adding an emitter follower (EF), you lower the impedance, so you can then precisely control it with a series resistor. The loss of signal is of no consequence. Make the stage gain 26dB if it worries you that much. :-)

Understood.

Yes. But not a very good approach, unless you're a high volume manufacturer desperate to save every part. Here, more is better.

Lucky break.

Exactly, thereby precisely setting Zo = 560 ohms to insure the filter would have its designer-intended frequency response.

Use the EF with say 2mA for Zo = 12 ohms (and with over 500mV filter-drive capability), and follow it with a 560-ohm resistor. That's 572 ohms, close enough! Or follow it with 549 ohms 1% to be right on the money.

Improve, yes, but get you where you want to be, no.

Loading the amplifier's high Zo directly with the filter is a dangerous game, creating distortion as you unwisely rob it of the excess loop gain it should be using to keep distortion down.

Emitter follower. Cathode follower (CF) without a filament. :-)

Like most things in electronic design there are trade offs. I wanted to be able to more fully understand the realtionship between Zo, the feedback ratio and operating current.

My thought was that it had to be be better than a 560 ohm resister in the drain circuit and amplifier stage with no feed back at all. The objective was to improve the signal handling capability and dynamic range without messing up the match by to much.

For the parameters with IC and N I need an equation that provides al least a ball park figure for Zo. Try as I may I can't see how to solve this.

Gain has its own problems of dynamic range with high Zo with a Vs of 10Volt as well as IP3 for the stage. Feedback has the advantage of added linearity reducing other undesired products of ajacent and large signals.

Sure but I'll bet IP3 and overall linearity are considerably worse.

I would be more tempted to run at 10mA ;-) and 2.5 ohms with RE of 470E. Then Rseries match of 560E. Should Zin/hFE not be added to this?

Sorry I have never been much of a fan of resistive matching for RF unless absolutely no other better way exists. One needs to achieve a good design balance for the full spectrum of requirements most times.

Retro fitting to existing circuits has it own problem of never enough space or remote mounting adding even more problems. But if the transformer feedback proved inferior I'll give it a try.

A perfect match is not the only requirement but one of a set involving noise, dynamic range, large signal handling, linearity.... It takes a considerable amount of time and often one has to think real hard to find a way of evaluating what you need to know. More frustrating is perhaps better because I know how but lack the equipment. For messing about there is no chance of recovering a large outlay on seldom used equipment.

But a sage with no feedback has no such protection or is the IC current drive to the resister match drain load sufficient to overcome the reflected impedences?

I am happy to reduce the gain down to 6 dB or whatever the noise figure is at the filter input plus 1 if needed because it is far easier to add a good low noise 50 ohm in/out stage at the antenna input ahead of the preselector or the preselector input which is 50E and 1st RF amp.

Ok but out of interest and because it has now become a mission I would still like to know how to calculate the approx Zo for the emitter feedback circuit given.

I am busy putting the final touches to ALC on my cheap and cheerful Chinese signal generator and that will allow me to more easily check the filter under both conditions. I gave up trying to control the tubes large output with junction fets or cathode degeneration and instead regulated the plate voltage. Pair of signal diodes + TL431 + 4N35 optocoupler and IRF830 as the series pass element feeding a 150E resister ahead of the smoothing capacitor. Rectified and regulated heater voltage was used to supply the TL431 and opto diode. The 4N35 CE shunts the gate and 15 volt protection zener driven by a 100K resister by the plate voltage connected to the drain. Crude but works well once the TL431 is stabilised. Then to modify my old AVO signal generator with 6J5 tube and figure out how to mount the stuff. ;-) A BF495 and J310 cascode works fine as a 6J5 replacement.

Regards and thanks for your adviced and insight to the problem.

Peter

Your design criteria seems to me to harken back to the days of expensive components and assembly costs, minimal parts count, and painful "optimization" (which, in many celebrated cases, absolutely did *not* work out). Not being disrespectful, but thankfully I left that design approach behind 38 years ago.

You're comparing apples and oranges: A common-source stage with a highly-linear predictable Zo and an uncontrolled-distortion, with a precisely-controlled low-distortion gain stage with a fragile "don't-tamper-with-me" Zout. That's how I see it. I prefer the second choice, because I can easily fix its problems and thereby achieve its low-distortion low-noise Nirvana.

IMHO, the low-distortion gain stage is the winner, hands down.

You're very welcome. I'm going to retire from the conversation, because your interests seem to move quite separately from mine, and I have too many pressing projects on the burner now. By all means, enjoy yourself. Let us know how your project progresses.